Features

Single power operation

Broadband: 4MHz

Low -dimensional voltage: 65 μV

The unit gain stable

High conversion conversion Rate: 4.0V/μs

Low noise: 3.9 nv/√Hz

Application

battery power supply instrument

Power control and protection

[ 123] Telecom

DAC output amplifier

ADC input buffer

General description

OP184

/

OP284 [123 ]/ OP484 is a single power supply, dual power supply and four -way single power supply, 4MHz bandwidth bandwidth, with rail pairing input and output. They ensure that they work within the voltage range of 3V to 36V (or ± 1.5V to ± 18V) and work under a voltage of 1.5V. These amplifiers are first -rate for single -power applications that require communication and precision DC performance. Combining bandwidth, low noise and accuracy, OP184/OP284/OP484 can be used for multiple applications, including filters and instruments.

Other applications of these amplifiers include portable telecommunications equipment, power control and protection, and amplifier or buffer as a wide output range sensor. The sensor of the rail -to -orbit input amplifier includes Hall effect, piezoelectric and resistor.

The ability of the input and output rail enables the designer to build a multi -stage filter in the single power system and maintain a high signal -to -noise ratio.

OP184/OP284/OP484 specified within the temperature range of the heat expansion (-40 ° C to+125 ° C). Single roots can be installed with 8 -line SOIC surface. There are 8 -line PDIP and SOIC surface installation packaging. The four -way OP484 has 14 -line PDIP and 14 -line narrow body SOIC packaging.

pin configuration

Simplified schematic diagram

Typical performance characteristics

] Application information Functional description OP184/OP284/OP484 is a precision single power supply and rail -to -rail transportation large device. For the portable instrument market, the OPX84 series equipment combines the characteristics of accuracy, broad band width and low noise to make it a need to makeThe best choice for the single power application of flow and precision DC performance. OP284 is very suitable for other low -power voltage applications include active filter, audio microphone front placing large, power control and telecommunications. In order to combine all these attributes with the input/output operation of the rail pairing, new circuit design technology is adopted. For example, FIG. 44 shows the simplified equivalent circuit used for input levels for OP184/OP284/OP484. It includes NPN differential pairs Q1 → Q2 and PNP differentials, Q3 → Q4, and run at the same time. The diode network D1 → diode network D2 is used to cut the applied differential input voltage to OP284, thereby protecting the input transistor from the damage to the avalanche. The input -level voltage gain is maintained at a lower input orbid. Two pairs of differential output voltage are connected to the second level of OP284, which is a composite folding level increase. It is also at the second increase level, and two of them are combined into a single -end output signal voltage for driving output level. A key problem in the input phase is the behavior of the input bias current in the input co -mode voltage range. The input bias current in OP284 is q1 → Q3 and Q2 → Q4 The albums of the base current. Due to this design method, the input bias current in OP284 not only shows different amplitude, but also shows different polarity. The best description of this effect is Figure 10. Therefore, it is important that the effective source impedance connected to the OP284 input terminal should be balanced to obtain the best DC and AC performance.

In order to achieve rail transfers, the OP284 output stage was designed with a unique topology structure to generate and absorb current. The circuit topology is shown in Figure 45. The output level is driven by the second -level gain level. Through the output -level signal pathway, it is reversed; that is, for the positive input signal, Q1 provides a base current driver for Q6 so that it can transmit (absorb) current. For negative input signals, the signal channel of Q1 → Q2 → D1 → Q4 → Q3 provides a base current driver for Q5 conduction (source) current. Both amplifiers provide output current until they are forced into the saturation state, and the saturation occurs when the negative power rail is about 20 millivolo and the positive electrode power rail about 100 millivolves.

Therefore, the saturated voltage of the output transistor sets the limitation of OP284 maximum output voltage swing. The output short -circuit current limit is determined by the maximum signal current that enters the Q1 base from the second gain level. Under short -circuit conditions, the input current is about 100 μA. The current of the transistor is about 200, and the short -circuit current limit is usually 20 mA. The output level also shows the voltage gain. This is achieved by using a public transmission polar amplifier. Therefore, the output -level voltage gain (therefore, the opening gain of the device) shows the dependencies of the total load resistance of the OP284 output place.

Enter over pressure protection

Like any semiconductor device, if the input voltage applied to the device exceeds any power voltage, the input over-voltage I-V characteristics of the device must be considered. When a voltage occurs, the amplifier may be damaged, depending on the size of the external voltage and the size of the fault current. Figure 46 shows the overvoltage I-V characteristics of OP284. This graphic is generated by a set of power pins connected to GND and a collector output drive that is connected to the input curve tracker.

As shown in Figure 46, internal P-N connected to OP284, when the input is 1.8 V and a negative 0.6 V higher than the respective power rail, the allowable current can be input from input from input Flow to power. As shown in the simplified equivalent circuit shown in Figure 44, the OP284 does not have any internal flow resistance; therefore, the fault current can rise rapidly to the destructive level.

The input current does not cause inherent damage to the device, provided that it is limited to 5 mAh or less. For OP284, once the input exceeds 0.6 V, the input current exceeds 5 mA. If this situation continues to exist, an external series resistor should be added at the cost of additional thermal noise. Figure 47 shows the typical non -conversion configuration of over -voltage protection amplifiers. The selection of the series resistance RS is as follows:

For example, the 1 kΩ resistor protection OP284 is not affected by the power supply up and down the power supply up and down The influence of 5V input signal. For other configurations of two inputs, each input should be used to protect each input to prevent abuse. Similarly, in order to ensure the best DC and AC performance, it is recommended to balance the level of power impedance.

The output phase reversed

Some operational amplifiers designed for the work of a single power supply. When the input is driven by the effective common modulus range, it will The output voltage phase reverses. Generally, for the single -power bilateral operational amplifier, the negative power source determines the lower limit of the scope of the co -mode. These devices are external clamping diode, anode grounding, and cathode input to prevent the input signal from offset the negative power supply (ie GND) of the device to prevent changes in the phase of the output voltage. The JFET input amplifier can also display the phase reversal. If so, a series of input resistance is usually required to prevent it.

As long as the input voltage does not exceed the power supply voltage, OP284 is not limited by a reasonable input voltage range. Although the output of the device does not change the phase, the large current can flow through the input protection diode, as shown in Figure 46. Therefore, the technology recommended in the input voltage protection section should be applied to applications with high possibilities of input voltage exceeding the power supply voltage.

The low -noise circuit design in the single power supply application

In the application of a single power supply, devices like OP284 expand the dynamic range of the application by using rail operations. In fact, OPXThe 84 series is the first product that combines single power, rail -to -track operation and low noise. It is the first device in the industry to display the input noise voltage spectrum density of less than 4nv/√Hz under 1KHz. It is also designed specifically for low noise and single power supply. Therefore, it is appropriate to discuss some discussions on the concept of circuit noise in single power applications.

Refer to the computing amplifier noise model circuit configuration in FIG. 48, the expression of the total equivalent input noise voltage of the source resistance level RS amplifier is shown below:

Among them:

RS u003d 2R is an effective or equivalent circuit source resistance.

(ENOA) 2 is an equivalent input noise voltage spectrum power (1HzbW) of the operation amplifier equivalent.

(INOA) 2 is an equivalent input noise current spectrum power (1HzbW) of the operation amplifier.

(ENR) 2 is source resistance thermal noise voltage (4KTR).

K u003d Bolitzman constant u003d 1.38 × 10–23 j/k.

T is the ambient temperature of the circuit, and the unit is Kelvins u003d 273.15+TA (° C).

As a design assistance, Figure 49 shows the total thermal noise of OP284's total input noise and the total thermal noise of the resistor in order to compare. Note that for the source resistance of less than 1kΩ, the equivalent input noise voltage of OP284 dominates.

Since the circuit signal -to -noise ratio is the key parameter of the final analysis, the noise characteristics of the circuit are usually represented by noise coefficient NF. The noise coefficient refers to the output signal -to -noise ratio of the circuit and the input signal -to -noise ratio. The expression of the circuit NF in DB as the unit, and the previously defined operation amplifier voltage and current noise parameters, from the following formula:

where:

[ 123] NF (DB) is the noise coefficient of the circuit, which is represented by DB.

RS is an effective or equivalent resistance provided to the amplifier.

(ENOA) 2 is OP284 noise voltage spectrum power (1 Hz BW).

(INOA) 2 is OP284 noise current spectrum power (1HzbW).

(ENRS) 2 is source resistance thermal noise voltage power u003d (4KTRS).

The noise coefficient of the circuit is easy to calculate because the signal level is not required to determine it in the application. However, many designers who use NF computing as the basis for obtaining the best signal -to -noise ratio believe that the low noise coefficient is equal to low noise. In fact, the opposite is the opposite, as shown in Figure 50. Here, the noise coefficient of OP284 indicates that the source resistanceThe level of the level. Please note that the minimum noise coefficient of OP284 appears at the source resistance level of 10 kΩ. However, FIG. 49 shows that the source of the source resistance is flat and OP284 generates about 14 nv/√Hz total equivalent circuit noise. In the application, the signal level is unchanged to maximize the circuit signal -to -noise ratio, which is not an option in low -voltage and single power applications.

Therefore, in the application of single power supply, in order to obtain the best circuit signal -to -noise ratio, it is recommended to choose an equivalent input noise voltage calculation amplifier, and select and keep low Source resistance level with consistent total circuit noise.

Over -speed recovery

The excess drive recovery time of the operation amplifier is the time required for the output voltage from the saturation state to its linear area. In the application that the amplifier must be quickly recovered after a large transient event, recovery time is important. The circuit shown in FIG. 51 is used to assess OP284 overload recovery time. OP284 recovers about 2 μs from positive saturation, and recovered from negative saturation by about 1 μs.

Single power supply, 3V meter amplifier

OP284's low noise, wide band wide and rail input/output operations make it a low power voltage application The ideal choice, as shown in Figure 52. The circuit uses a classic dual -transportation instrument to set top topology and four resistors to set up gain. The transmission equation of the circuit is the same as the transmission equation of non -rotary large -handed largers. The resistor R2 and the resistor R3 should be closely matched, and the resistor (R1+P1) and the resistor R4 should be ensured to ensure good co -mode inhibitory performance. The resistance network is applied to the R2 and R3 circuits because they show the relative tolerance matching required for good performance. The matching network also shows a tight relative resistance temperature coefficient to obtain good circuit temperature stability. Fine -tuning potential P1 is used to optimize DC CMR adjustments, and C1 is used to optimize AC CMR. The circuit value is as shown in the figure that the CMR CMR is better than 80 decibels in the frequency range of 20 Hertz to 20 kilo. The RTI (referred to input) in the 0.1 Hz to 10 Hz band is very low, with 0.45 μV P-P. The resistor RP1 and resistor RP2 are used to protect OP284 input and prevent input from input overgrigger abuse. Capacitor C2 may include in a restricted circuit bandwidth, so broadband noise in sensitive applications. The value of the capacitor should be adjusted according to the closed bandwidth required by the circuit. The time constant of R4 to C2 is equal to

2.5 V reference voltage from 3 V power supply

In many single power supply applications, 2.5 V references often need Voltage. Many commercial single -piece 2.5 V reference voltage requires at least 4 V's minimum working power supply. When the minimum working power supply voltage is 3 V, the problem is more serious. The circuit shown in FIG. 53 is an example of a 2.5 V reference voltage.Power from a single 3 V power supply. The circuit uses the input/output voltage range between OP284 rails to amplify the 1.235V output of AD589 to 2.5V.

low TCVO at 1.5μV/° C in OP284 helps maintain the output voltage temperature coefficient controlled by the temperature coefficient controlled by R2 and R3. In this circuit with a 100 PPM/° C TCR resistor, the temperature coefficient of the output voltage is 200 PPM/° C. It is recommended to use a low temperature resistor to obtain more accurate ultra -temperature performance.

One indicator of measurement of voltage benchmark performance is the ability it to recover from the sudden change of the load current. When the steady state load current is 1mA, the circuit is restored to 0.01%of the programming output voltage within 1.5 μs, and the total change of the load current is ± 1mA.

Only 5V, 12 -bit DAC rail swing

OP284 is very suitable for use with CMOS DAC to generate digital control voltage with a wide output range Essence FIG. 54 shows the DAC8043 used with AD589 to generate voltage output from 0 V to 1.23 V. DAC actually works in the voltage switch mode, where the benchmark is connected to the current output iOUT, and the output voltage is taken from the VREF pin. Contrary to the traditional current output mode, this topology is essentially irreversible and is not available in single power applications.

In this application, OP284 provides two features. First of all, it cushioned the high output impedance of the VREF pin of DAC, about 10kΩ. The computing amplifier provides low impedance output to drive any follow -up circuit.

Second, the operational amplifier amplifier output signal is provided to provide rail -to -orbit output swing. In this special case, the gain is set to 4.1, so that the circuit generates 5V output when the DAC output is full standard. If you need other output voltage range, such as 0V ≤Vout≤4.095V, you can easily change the gain by adjusting the values u200bu200bof R2 and R3.

High -voltage side current monitor

In the design of the power control circuit, a large number of design work is concentrated in ensuring the long -term reliability of the augmented crystal tube under the extensive load current conditions. Therefore, in these designs, monitoring and restricting equipment power consumption are the most important. The circuit shown in FIG. 55 is an example of a 3V single -power high -voltage side current monitor. The monitor can be composed of a voltage regulator with a foldable flow limit function or a large current power supply with a pry -pole protection. The design uses the orbit input voltage range of the OP284 to detect the voltage drop on the 0.1Ω current distributor. The P -channel MOSFET used as a feedback element converts the differential input voltage of the computing amplifier to current. This currentApply to R2 to generate a voltage, which is a linear representation of the load current. The transmission equation of the current monitor is shown below:

For the displayed component value, the transmission characteristics of the monitor output are 2.5 v/a.

Capacity load driving capacity

OP284 has excellent capacitor load driving capabilities. It can drive up to 1Nf at most, as shown in Figure 28. Even if the device is stable, the capacitance load will not cause losses in bandwidth. For loads greater than 2NF, the bandwidth is reduced to less than 1MHz. The buffer network at the output end does not increase the bandwidth, but it can indeed significantly reduce the over -adjustment of a given capacitance load.

The buffer consists of a series R-C network (RS, CS), as shown in Figure 56, connecting from the output end of the device to the ground. The network work with the load capacitor CL parallel to provide the necessary phase lag compensation. The value of resistance and capacitors is best determined according to experience.

The first step is to determine the value of the resistance RS. A good start value is 100Ω (usually, the best value is less than 100Ω). This value is reduced until the small signal transient response is optimized. Next, measure CS; 10μF is a good starting point. This value is reduced to the minimum value of acceptable performance (usually 1 μF). For the case of 10NF load capacitors on OP284, the best buffer network is 20Ω series 1 μF. The benefits are obvious, as shown in the oscilloscope in Figure 57. The top record is collected under the 1 NF load, and the bottom record is collected under the position of 50Ω and 100 NF buffer networks. The quantity of overnaming and ringing is an amazing reduction. Table 6 shows some buffer network examples for large load capacitors.

The current low voltage difference variable regulator

Many circuits need a stable, adjustable voltage, the voltage is relatively close to Unexpected input source. This low -voltage type regulator is easy to implement with a rail -to -orbit output op amp, such as OP284, because the width output width allows it to drive the low saturation voltage through the device. In addition, when the operational amplifier also adopts rail -to -track input characteristics, it is particularly useful because this factor allows it to perform high -side current influenza to achieve positive rail current restrictions. A typical example is a system power supply from 3V to 9V or any voltage that requires low voltage drops to improve power efficiency. This 4.5V example works on the nominal power supply of 5V, and the level of the worst case drops to 4.6V or less. Digital 58 shows such a regulator settings, using an OP284 plus a low RDS (on) P channel MOSFET through the device. Part of the low voltage difference between the circuit is provided by Q1, its rated value is 0.11Ω, and the grid drives electricalThe pressure is only 2.7 V. This relatively lower gate driver threshold allows the regulator to run on a power supply as low as 3V without affecting the overall performance.

The main voltage control loop operation of the circuit is provided by half of the OP284. The terminal voltage of this voltage control amplifier is 192. Then adjust the output voltage:

In this example, when VOUT is 4.5 V and VOUT2 u003d 2.5 V, the U1B gain is 1.8 times, so the ratio of R3 and R2 chooses R3 and R2 It is 1.2: 1 or 10.0 kΩ: 8.06 kΩ (using the closest 1%value). Note that for the lowest VOUT DC error, the R2 | R3 should remain equal to R1 (as shown in this case), and the R2 to R3 resistor should be a stable metal membrane type with close tolerance. The table in Figure 58 summarizes some commonly used voltage R1 to R3 values. However, please note that under normal circumstances, the output can be between anywhere between VOUT2 and 12 volts of the maximum rated value Q1.

The low voltage saturation characteristics of Q1 are the key part of the low voltage difference, and another component is a low -current influenza comparative threshold with good DC accuracy. Here, this is provided by the current detection amplifier U1A. It is provided by 20 millivolt volt voltage from 1.235V, AD589 reference diode D2 and R8 to R8 -to -R8 -point frequency frequency. When the output current and the product of the RS value match the voltage threshold, the current control loop is activated, and U1A drives the Q1 gate through D1. This will cause the entire circuit operation to enter the current mode control, and the current limit Ilimit is defined as:

It is obvious that keeping this comparison voltage is very small. It is desirable because it has become an important part of the entire voltage loss. Here, the typical offset of the 20 millivolo reference voltage is higher than the OP284, but the percentage of VOUT is still quite low (u0026 lt; 0.5%). When the limiter adapts to other Ilimit levels, the sensor RS should be adjusted with R7 to R8 to keep the threshold voltage between 20 MV and 50 MV.

Good circuit performance. For the 4.5 V output version, the DC output changes measured under 225 mA load changes are about a few slightly, and the voltage drop voltage at the same current level is about 30 millivolves. As shown in the figure, the current is limited to 400 mAh, and the allowed circuit to be used at a level of 300 mAh or higher. The Q1 device can actually support the current of several ampels. The actual rated current considers the loss of 2.5W and 25 ° C of the SOIC-8 device. Because the short -circuit current of 400 mAh when the input level is 5V will cause 2 W loss in Q1, other input conditions should be considered according to the potential overheating of Q1. Of course, if Q1 uses higher power devices, this circuit can support the losses of dozens of ampels.And the higher VOUT level that has been mentioned.

The circuit shown in

can be used as a standard low -voltage differential regulator, or it can also be used for opening/control. Use optional logic control signal VC to drive the No. 3 pin of U1, and the output is switched between the switch. Please note that when the output in this circuit is disconnected, it is still in the activation state (that is, not the road). This is because the shutdown state only reduces the input voltage to R1, so that the U1A/U1B amplifier and Q1 are still effective.

When the opening/level control is used, the resistor R10 should be used with U1 to accelerate the opening/control, and the circuit output is stable to the nominal zero voltage. Component D3 and component R11 also help the C2 to help accelerate the switch conversion. The switch conversion time is less than 1ms, and the switch conversion time is long, but less than 10ms.

3 V, 50 Hz/60 Hz Affiliated wave filter, with fake ground

In a single power supply system, it is best to use a fake ground bias scheme. The circuit using this method is shown in Figure 59. In this circuit, the fake ground circuit causes the source of the source to be biased, and the source trap filter is used to inhibit the 50 Hz/60 Hz power cord interference in the guardians of portable patients.

Paper filter is usually used to inhibit the frequency interference of the power line. These interference usually cover up low -frequency physiological signals, such as heart rate, blood pressure reading, electro -electrocardiography, and electrocardiogram. This trap filter effectively inhibits 60 Hz picks of the filter Q of 0.75. Use a 3.16 kΩ resistor to replace the 2.67 kΩ resistor in the TWIN-T (R1 to R5), and configure the active filter to inhibit 50 Hz interference.

The amplifier A3 is the core of a fake ground bias circuit. The voltage generated by its buffer in R9 and R10 is a reference for source trap filters. Because OP484 has the scope of the rail -to -orbit input, the 3V power supply is selected symmetrically at R9 and R10. A circular compensation scheme was used around OP484, allowing the operational amplifier to drive C6, a 1μF capacitor without oscillation. C6 maintains low impedance exchange ground within the operating frequency range of the filter.

The filter part adopts OP484 with a dual T structure, and its frequency selectivity is very sensitive to the relative matching of the capacitance and resistance in the dual T section. Polycardia film is the first choice of capacitors. The relative matching of the capacitor and resistance determines the pseudonym of the filter. Use 1%resistance and 5%capacitor can produce satisfactory results.

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